The present invention relates to a switching power supply circuit to be provided as a power supply for various electronic apparatus.
Switching power supply circuits employing switching converters such for example as flyback converters and forward converters are widely known. These switching converters form a rectangular waveform in switching operation, and therefore there is a limit to suppression of switching noise. It is also known that because of their operating characteristics, there is a limit to improvement of power conversion efficiency.
Hence, various switching power supply circuits employing resonance type converters have been proposed. A resonance type converter makes it possible to readily obtain high power conversion efficiency, and to achieve low noise because the resonance type converter forms a sinusoidal waveform in switching operation. The resonance type converter has another advantage of being able to be formed by a relatively small number of parts.
FIG. 8 is a circuit diagram showing an example of a related art switching power supply circuit. The fundamental configuration of the power supply circuit shown in the figure has a voltage resonance type converter as a primary-side switching converter.
The power supply circuit shown in the figure generates a rectified and smoothed voltage Ei whose level is equal to that of an alternating input voltage VAC from a commercial alternating-current power by a bridge rectifier circuit Di and a smoothing capacitor Ci.
The voltage resonance type converter for interrupting the rectified and smoothed voltage Ei (direct-current input voltage) inputted thereto includes a switching device Q1 and employs a single-ended system. The voltage resonance type converter employs a self-excited driving method. In this case, a high voltage bipolar transistor (Bipolar Junction Transistor) is selected as the switching device Q1 forming the voltage resonance type converter. A primary-side parallel resonant capacitor Cr is connected in parallel with a collector and an emitter of the switching device Q1. A clamp diode DD is connected between a base and the emitter of the switching device Q1. The parallel resonant capacitor Cr forms a primary-side parallel resonant circuit in conjunction with leakage inductance L1 obtained in a primary winding N1 of an isolating converter transformer PIT, whereby operation of the voltage resonance type converter is obtained.
A self-oscillation driving circuit formed by a driving winding NB, a resonant capacitor CB, and a base current limiting resistance RB is connected to the base of the switching device Q1. The switching device Q1 is driven for switching operation by being supplied with a base current based on an oscillating signal generated by the self-oscillation driving circuit. At the start of power supply, the switching device Q1 is started by a starting current flowing from the rectified and smoothed voltage Ei line to the base of the switching device Q1 via a starting resistance Rs.
In this case, in addition to the clamp diode DD connected between the base and emitter of the switching device Q1, a clamp diode DD1 is connected between the collector and emitter of the switching device Q1.
An orthogonal type control transformer PRT is formed by winging a control winding NC in a winding direction orthogonal to a current detecting winding ND and a driving winding NB. The orthogonal type control transformer PRT is provided to control switching frequency of the primary-side voltage resonance type converter. The structure of the orthogonal type control transformer PRT is a cubic core formed by connecting two table-shaped cores each having four magnetic legs with each other at ends of the magnetic legs. The resonance current detecting winding ND and the driving winding NB are wound around two given magnetic legs of the cubic core in the same winding direction, and the control winding NC is wound around magnetic legs in a direction orthogonal to the resonance current detecting winding ND and the driving winding NB.
An isolating converter transformer PIT (Power Isolation Transformer) is provided to transmit the switching output of the switching converter obtained on the primary side to the secondary side of the switching power supply circuit. The isolating converter transformer PIT is formed by winding the primary winding N1 and secondary winding N2 of the isolating converter transformer PIT around an Exe2x80x94E-shaped core in a state of being divided from each other. Also, a gap G is formed in a central magnetic leg of the Exe2x80x94E-shaped core. Thus, loose coupling at a desired coupling coefficient is obtained, and accordingly a saturated state is not readily obtained.
The primary winding N1 of the isolating converter transformer PIT is connected between the line of the direct-current input voltage and the collector of the switching device Q1. The switching device Q1 performs switching operation on the direct-current input voltage. Thus, the primary winding N1 is supplied with the switching output of the switching device Q1, and thereby generates an alternating voltage having a cycle corresponding to the switching frequency of the switching device Q1.
The alternating voltage induced by the primary winding N1 is generated in the secondary winding N2 on the secondary side of the isolating converter transformer PIT. In this case, a secondary-side parallel resonant capacitor C2 is connected in parallel with the secondary winding N2. Thereby, leakage inductance L2 of the secondary winding N2 and capacitance of the secondary-side parallel resonant capacitor C2 form a parallel resonant circuit. The parallel resonant circuit converts the alternating voltage induced in the secondary winding N2 into a resonance voltage. Thus, a voltage resonance operation is obtained on the secondary side.
Thus, the power supply circuit is provided with the parallel resonant circuit to convert switching operation into voltage resonance type operation on the primary side and the parallel resonant circuit to provide voltage resonance operation on the secondary side. The switching converter provided with such resonant circuits on the primary side and the secondary side is referred to as a xe2x80x9ccomplex resonance type switching converter.xe2x80x9d
On the secondary side of the isolating converter transformer PIT in this case, an anode of a rectifier diode D01 is connected to a winding end point of the secondary winding N2, and a cathode of the rectifier diode D01 is connected to a positive electrode terminal of a smoothing capacitor C01, thereby forming a half-wave rectifier circuit. The half-wave rectifier circuit provides a secondary-side direct-current output voltage E01 across the smoothing capacitor C01.
In this case, the secondary winding N2 is provided with a tap, and a half-wave rectifier circuit formed by a rectifier diode D02 and a smoothing capacitor C02 is connected to the tap output, as shown in the figure. The half-wave rectifier circuit provides a secondary-side direct-current output voltage E02 that is lower than the secondary-side direct-current output voltage E01. Incidentally, the secondary-side direct-current output voltage E01 is 135 V, and the secondary-side direct-current output voltage E02 is 15 V, for example.
The secondary-side direct-current output voltages E01 and E02 are each supplied to a required load circuit. The secondary-side direct-current output voltage E01 is outputted from a branch point as a detection voltage for a control circuit 1, and the secondary-side direct-current output voltage E02 is outputted from a branch point as operating power for the control circuit 1.
The control circuit 1 supplies the control winding NC of the orthogonal type control transformer PRT with a direct current that is variably changed according to the level of the secondary-side direct-current output voltage E01 as a control current. In response to the change in the level of the control current flowing through the control winding NC, the orthogonal type control transformer PRT variably controls the inductance LB of the driving winding NB. This results in a change in resonance frequency of the resonant circuit formed by the driving winding NB and the resonant capacitor CB in the self-oscillation driving circuit. The switching frequency of the switching device Q1 is thereby variably controlled. The switching frequency of the switching device Q1 is thus changed to control the secondary-side direct-current output voltage at a constant level. Thus, the power supply is stabilized.
Assuming that a relation of Po1 greater than  greater than Po2 holds for load powers Po1 and Po2 of the secondary-side direct-current output voltages E01 and E02, the circuit shown in FIG. 8 forms operating waveforms as shown in FIGS. 9A to 9C under a condition of a maximum load power.
FIG. 9A shows a parallel resonance voltage V1 obtained across the primary-side parallel resonant capacitor Cr. As shown in the figure, the parallel resonance voltage V1 is at a zero level during a period TON during which the switching device Q1 is turned on, and forms a sinusoidal pulse waveform during a period TOFF during which the switching device Q1 is turned off.
As shown in FIG. 9B, a switching current Icp flowing to a parallel circuit of the switching device Q1 and the clamp diode DD1 is at a zero level during the period TOFF. During the period TON, the switching current Icp forms a waveform such that a damper current in a direction of negative polarity first flows and thereafter a collector current in a direction of positive polarity flows. As shown in FIG. 9C, during the period TON, a damper current ID1 flowing through the clamp diode DD1 forms a waveform of positive polarity in the damper current period of the switching current Icp of FIG. 9B.
As is understood from these waveforms, the parallel resonance voltage V1 shown in FIG. 9A is obtained during only the period TOFF, while the switching current Icp and the damper current ID1 shown in FIGS. 9B and 9C, respectively, are obtained only during the period TON. Hence, normal xe2x80x9cZVS (Zero Voltage Switching) operationxe2x80x9d is obtained.
When the power supply circuit shown in FIG. 8 is to be incorporated in a television receiver, for example, supply voltages having different levels are supplied as secondary-side direct-current output voltages to various circuit units. Therefore, it is necessary to generate and output not only the secondary-side direct-current output voltages E01 and E02 but also other secondary-side direct-current output voltages.
A configuration on the secondary side of the isolating converter transformer PIT in this case is shown in FIG. 10. In the figure, the same parts as in FIG. 8 are identified by the same reference numerals, and their description will be omitted.
The figure shows that five direct-current voltages, that is, secondary-side direct-current output voltages E01 to E05 are generated.
The secondary-side direct-current output voltage E01 is obtained by the same circuit configuration as in FIG. 8. Specifically, the secondary-side direct-current output voltage E01 is obtained by the half-wave rectifier circuit formed by the rectifier diode D01 and the smoothing capacitor C01 and connected to the secondary winding N2.
The tap output of the secondary winding N2 is connected with the rectifier diode D02 and the smoothing capacitor C02, and further the smoothing capacitor C02 and a smoothing capacitor C03 to be described later are connected in series with each other. Thus, a current rectifying path is formed by the rectifier diode D02 and a series connection circuit of the smoothing capacitors C02 and C03, whereby a secondary-side direct-current output voltage E02 of 15 V, for example, is obtained across the smoothing capacitors C02 and C03.
Further, a center tap is provided between the tap output for the secondary-side direct-current output voltage E02 and the winding start point of the secondary winding N2, and a half-wave rectifier circuit of a rectifier diode D03 and a smoothing capacitor C03 is provided for the center tap, as shown in the figure, whereby a secondary-side direct-current output voltage E03 of 7.5 V, for example, is obtained.
In this case, a step-up winding N3 having a given number of turns is formed by winding an additional wire from the winding end point side of the secondary winding N2. The step-up winding N3 is connected with a rectifier diode D04 and a smoothing capacitor C04 as shown in the figure. A negative electrode terminal of the smoothing capacitor C04 is connected to a positive electrode terminal of the smoothing capacitor C01, thereby forming a series connection circuit of the smoothing capacitors C04 and C01. Thus, a current rectifying path is formed by the rectifier diode D04 and the series connection circuit of the smoothing capacitors C04 and C01, whereby a secondary-side direct-current output voltage E04 of 200 V is obtained across the smoothing capacitors C04 and C01.
Furthermore, in this case, an independent secondary winding N4 is wound on the secondary side of the isolating converter transformer PIT independently of the secondary winding N2 and the step-up winding N3. As shown in the figure, a half-wave rectifier circuit of a rectifier diode D05 and a smoothing capacitor C05 is provided for the independent secondary winding N4, whereby a secondary-side direct-current output voltage E05 of 24 V, for example, is obtained.
When the power supply circuit employs the configuration for generating and outputting a relatively large number of secondary-side direct-current output voltages as shown in FIG. 10, power obtained from the secondary-side winding N2 accompanied by the resonant capacitor C2 is less than power obtained from another secondary-side winding N4, the same parts as those of FIGS. 9A to 9C exhibit waveforms as shown in FIGS. 11A to 11C. As is shown by the waveforms of the switching current Icp and the damper current ID1 of FIGS. 11B and 11C, the absolute value levels of the currents flowing during the damper period are lower than those shown in FIGS. 9B and 9C. This means that a margin of a ZVS operation range is reduced.
When a load power of 115 V outputted from E01 is decreased in this condition, for example, the waveforms shown in FIGS. 11A to 11C are changed into states as shown in FIGS. 11D to 11F, respectively.
Specifically, as shown in FIG. 11F, the damper current ID1 that should flow through the clamp diode DD1 does not flow. As shown in FIGS. 11D and 11E, the parallel resonance voltage V1 and the switching current Icp both form abnormal operating waveforms. Thus, normal ZVS operation is not performed. Such abnormal operation significantly increases power loss, and an increase in the temperature of heat generated by the power loss may result in failure of devices such as the switching device Q1, for example.
As a measure against this, the resonance frequency of the secondary-side parallel resonant circuit may be set low by increasing the capacitance of the secondary-side parallel resonant capacitor C2. In this case, however, current flowing through the secondary winding N2 and the secondary-side parallel resonant capacitor C2 is increased, and thus the maximum load power that can be handled is decreased.
It is possible to secure the maximum load power when a turns ratio between the primary side and the secondary side is made larger by increasing the number of turns of the windings wound on the secondary side of the isolating converter transformer PIT or decreasing the number of turns of the primary winding N1, for example. In this case, however, power conversion efficiency is lowered, and power loss in the isolating converter transformer PIT and the switching device Q1 is increased, which results in a large amount of heat generated by the power loss.
In view of the above problems, a switching power supply circuit according to the present invention is configured as follows.
To achieve the above object, according to a first aspect of the present invention, there is provided a switching power supply circuit, including: switching means formed by including a switching device driven by switching driving for performing switching operation on a direct-current input voltage inputted thereto; a first isolating converter transformer for transmitting an output of the switching means obtained on a primary side of the first isolating converter transformer to a secondary side of the first isolating converter transformer; a primary-side parallel resonant circuit formed by a primary-side winding included in the first isolating converter transformer and a primary-side parallel resonant capacitor for converting operation of the switching means into voltage resonance type operation; and an inductance connected in parallel with the primary-side winding.
According to a second aspect of the present invention, there is provided a switching power supply circuit, including: a secondary-side resonant circuit formed by connecting a secondary-side resonant capacitor in parallel with a first secondary-side winding of the plurality of secondary-side windings; a first direct-current output voltage generating means for providing a first secondary-side direct-current output voltage by performing rectifying operation on an alternating voltage obtained in the first secondary-side winding and inputted to the first direct-current output voltage generating means; and a second direct-current output voltage generating means for providing a second secondary-side direct-current output voltage by performing rectifying operation on an alternating voltage obtained in another secondary-side winding than the first secondary-side winding and inputted to the second direct-current output voltage generating means, and for supplying more load power than the first direct-current output voltage generating means.
The switching power supply circuit has a configuration of a so-called complex resonance type switching converter, which is provided with the primary-side parallel resonant circuit for a voltage resonance type converter provided on the primary side and the secondary-side resonant circuit formed by the secondary-side resonant capacitor and the secondary-side winding on the secondary side.
A constant-voltage method in this case is a switching frequency control method that changes the switching frequency of the switching device.
With an inductor connected in parallel with the primary winding of the isolating converter transformer in such a configuration, the current flowing to the switching device has a normal waveform regardless of the level of load power. Therefore, it is possible to secure a margin of a ZVS operation range.
Thus, the connection of the inductance component eliminates the need to increase the capacitance of the secondary-side parallel resonant capacitor for normal ZVS operation when the load power is decreased. This results in a reduction of input power and an improvement in power conversion efficiency at a maximum load power in particular. In addition, as a result of this, generation of heat in the isolating converter transformer and the switching device forming the voltage resonance type converter is effectively controlled.